FIG. 1 shows the concept of a basic communication system 8 including a transmitter 12, a transmission medium 14 (such as a cable or wire) which is corrupted by noise 16, and a receiver 18. In serial digital data communications, the input signal 10 consists of an input pulse train or sequence. The input signal 10 is attenuated and distorted by the medium 14, through which it is transmitted, before a received signal 17 arrives at the receiver 18 which after processing the signal 17 provides the output signal 20. Distortion is caused by variable delay (dispersion) and variable attenuation of high frequency components. This distortion results in pulse spreading and consequential interference between neighbouring pulses known as ISI (intersymbol interference).
As shown in FIG. 1A, receiver 18 may typically include an automatic or adaptive equalizer 60 to offset the undesirable frequency effects of the cable (or other transmission medium), a DC (direct current) restorer 62 to restore or regenerate the DC component of the transmitted input, and an automatic gain control circuit 64 which provides the necessary gain for the equalizer 60, as explained below. The adaptive aspect of the cable equalizer is particularly useful, for example, where one receiver is capable of receiving several different signals transmitted from different locations and over cables having different lengths. FIG. 1B illustrates a communications system wherein a receiver 18 receives signals from a number of different transmitters (12-1, 12-2, 12-3, and 12-4) that respectively transmit over cables (14-1, 14-2, 14-3, and 14-4) which are of different lengths. An automatic cable equalizer in the receiver 18 should be able to equalize signals which have been transmitted over any cable length between some minimum length (e.g. zero length) and some maximum length.
Theoretically, an equalizer should have a frequency characteristic that is the inverse of the transmission medium and which restores high frequency components and eliminates dispersion. In practice however, this also increases noise at the receiver by increasing the noise bandwidth and boosting high frequency noise components. As is well known in the art, the loss over a cable (such as a co-axial cable) of length L may be approximated in frequency domain terms by: EQU L(j{character pullout})=e.sup.-AL(j{character pullout}).sup..sup.1/2 ,{character pullout}=2.pi..function.
where A is a constant. As is common practice and to facilitate understanding, the analysis of equalizer functionality is carried out in the frequency domain. Note that the function L(j{character pullout}) if expanded and expressed in the form of a numerator polynomial divided by denominator polynomial has an infinite number of poles and zeros. As a result, and as is further well known in the art, in a typical implementation of an automatic cable equalizer, the inverse cable loss function is approximated as: EQU G(j{character pullout})=1+Kf(j{character pullout}) PA1 where K is a control variable which varies depending on the length of the cable over which the signal was transmitted from zero at the minimum cable length to unity (or some other constant) at the maximum cable length. The equalizer function circuitry 22 is illustrated in FIG. 2 where the circuitry for providing the variable gain K is shown at 24, the circuitry which realizes the function f(j{character pullout}) is shown at 26, and the summing function is shown at 28. When the amplitude of the transmitted signal is a standard amplitude which is known, the amount by which the amplitude of the received signal (see below) has been attenuated may be used to provide an appropriate value for the gain K 25 (and correspondingly indicate the length of the cable over which the received signal was transmitted). As will be explained below, this may be obtained, via an AGC system and a DC restorer.
The poles and zeros of the function f(j{character pullout}) are chosen so that 1+f(j{character pullout}) provides a good approximation to the inverse cable loss L(j{character pullout}) at the maximum cable length. FIG. 2A illustrates a possible implementation of a circuit which may achieve an f(j{character pullout}) transfer function. Note that in FIG. 2A the f.sub.in and f.sub.out signals, which are respectively the input and output of the f(j{character pullout}) circuit, are shown as differential signals whereas in FIG. 2 these signals are shown as single-ended. Referring to FIG. 2A, transistors 74 and 76 form a differential pair whose emitter terminals are connected through an impedance network 78 (each emitter terminal is also connected to a reference through current sources 80 and 82 respectively). The impedance network typically comprises a plurality of resistor-capacitor circuits cascaded together in parallel. The values of the resistor and capacitor components define the poles and zeros of f(j{character pullout}). The collectors of transistors 74 and 76 are coupled to Vcc through resistors 70 and 72 respectively. The input to f(j{character pullout}) is applied between the base terminals of transistors 74 and 76, and the output of f(j{character pullout}) is taken between the collector terminals of 74 and 76.
The equalization approach illustrated in FIG. 2 is, however, subject to several drawbacks. First, since the best approximation to the desired inverse cable loss response occurs at the extreme values of the control variable K, i.e. when K=0 (corresponding to the minimum cable length) and when K=1 (corresponding to the maximum cable length), the accuracy of the approximation deteriorates for intermediate values of K (corresponding to intermediate cable lengths). As the accuracy of the approximation worsens, the resulting errors cause increased jitter in the recovered data.
Second, the above approach is overly susceptible to noise associated with the f(j{character pullout}) function. Typically, the function f(j{character pullout}) can provide a gain of more than 40 dB at a frequency of 200 MHz. As shown in FIG. 2, to prevent overload of the f(j{character pullout}) function by the larger input levels associated with short cable lengths, the circuitry for the gain control function K 24 must be physically placed ahead or in front of the circuitry which realizes the f(j{character pullout}) function 26. As a result, the noise associated with the function f(j{character pullout}) is never attenuated and is always present at the output, irrespective of the value of K. Again, this causes an increase in jitter, particularly for lower values of K.
Third, the function G(j{character pullout}) is also chosen to delay high frequency signals in an inverse manner to the dispersion characteristic of the cable. When K is varied, the delay through the equalizer is also varied. Therefore when K varies in an undesirable manner, for example due to the presence of noise on the K controlling signal 25, the resulting delay modulation further contributes to jitter.
In addition, ideally a cable equalizer capable of multi-standards operation should be able to trade cable length for data rate as cable length is varied (for e.g., 800 Mbits/second at 100 metres, 200 Mbits/seconds at 400 metres). To minimize noise and ensure stability, the bandwidth of the function G(j{character pullout}) should also vary inversely with cable length. In practice, however, adding circuitry for realizing a variable bandwidth function to the equalizer of FIG. 2 results in increased circuit noise and delay modulation, and therefore jitter.
The above described problems render the cable equalizer of FIG. 2 overly susceptible to producing jitter. This prior art cable equalizer is also unsuitable for multi-standards use since standards with higher data rates, and consequentially shorter critical or maximum cable lengths, fall into the non-optimal intermediate operating region and because of the increased jitter levels associated with adding circuitry for providing variable bandwidth.
As already mentioned, the receiver 18 also typically includes a DC restorer to restore the DC component of the input pulse train and thereby eliminate baseline wander. Such a DC restorer may be a clamping DC restorer or a DC restorer based on the principle of quantized feedback (QFB). Both clamping and quantized feedback restorer circuits are described in detail in U.S. Pat. No. 5,426,389, the description of said patent being incorporated herein by this reference. FIG. 3 shows a standard quantized feedback (QFB) DC restorer 100 comprising a QFB comparator 150, positive feedback resistor 154, and input AC (alternating current) coupling capacitor 152. The DATA IN signal is coupled to the positive input terminal of comparator 150 through capacitor 152 and to the output of the comparator, i.e. the DATA OUT signal, through resistor 154. The negative input terminal of comparator 150 is connected to a reference voltage V.sub.ref. Because the restorer of FIG. 3 is configured with positive feedback, it has a bistable voltage transfer characteristic with hysteresis as illustrated in FIG. 3A. Referring to FIG. 3A, if the DATA OUT signal is low and the DATA IN signal is increasing in magnitude, the DATA OUT signal remains low until the DATA IN signal passes an upper threshold V.sub.H. Subsequently the DATA OUT signal remains high until the DATA IN signal decreases below a lower threshold V.sub.L. The values of V.sub.H and V.sub.L will depend on the values of resistor 154 and of capacitor 152.
The receiver 18 also typically includes an AGC circuit or an automatic control circuit which, in response to an error signal provided by a DC restorer circuit, may be used to control the K controlling signal 25. FIG. 4 shows a typical AGC system with a QFB DC restorer, such as that illustrated in FIG. 3. (Note that the AGC system 102 may include an automatic equalizer (not shown in FIG. 4) which not only adjusts the gain but also the frequency characteristic of the DATA IN signal, while, in other applications, the AGC system 102 may only provide a variable gain function to the DATA IN signal.) Although DC restorers using QFB are capable of low edge jitter performance (i.e minimizing spurious or random signal variations during data transitions), this requires controlling the amplitude of the input signal, DATA IN, to be very nearly equal to the output quantization level (in a sense, the quantization error must be minimized), since, as may be seen from FIG. 3, the DATA OUT voltage level, once established, will tend to follow the DATA IN voltage level. Generally, an automatic gain control (AGC) circuit automatically changes the gain or amplification of a receiver to maintain the desired output signal, or its amplitude, essentially constant despite variations in input signal strength. As shown in FIG. 4, when the input signal amplitude is controlled by an AGC function 102, as for example in the case of a line equalizer, the amplitude of the DC restorer input signal as detected by 104 is typically compared to the amplitude of the quantized signal as detected by 106 using a summer 108, and the difference, which is the quantization or AGC error signal 110, can be used to control the gain provided by the AGC circuitry 102.
However, the frequency spectrum of the quantized signal 114 and the controlled signal 112 generally differ at very low frequencies and very high frequencies since the transmitted or controlled signal 112 is AC coupled and band limited. As a result, there is a loss in the low frequency and high frequency components of the controlled signal 112 compared to the quantized signal 114 which has these components essentially reconstituted. The additional energy in the quantized reference signal 114 results in undesirably higher levels being produced by the amplitude detector 106 as compared to those produced by amplitude detector 104. This results in increased edge jitter in the output of the restorer.
In addition, a drawback of QFB DC restorers using positive feedback, such as that illustrated in FIG. 3, is that there is a 50% probability that the quantized output will be at the wrong level at the onset of data transmission. As a result, the comparator circuit may "latch-up" when data transmission first begins and operational failure of the DC restorer circuit may result, unless additional circuitry is employed to prevent such a situation. Typical solutions for avoiding this latch-up problem involve AC coupling the DATA OUT signal. However, the AC coupled output places a lower limit on the data rate and results in a deterioration of the system noise margin during periods when no data transitions occur. In addition, while clamping DC restorers are not susceptible to latch-up problems, they exhibit edge jitter performance which is inferior to QFB comparators.